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AN1330 Application note
Designing with the L5970D 1A high efficiency DC-DC converter
Introduction
The L5970D is a step-down monolithic power switching regulator capable of delivering up to 1 A at output voltages from 1.235 V to 35 V. The operating input voltage ranges from 4.4 V to 36 V. It has been designed using BCDV technology and the power switching element is implemented through a P-channel DMOS transistor. It does not require a bootstrap capacitor, and the duty cycle can range up to 100%. An internal oscillator fixes the switching frequency at 250 kHz. This minimizes the LC output filter. A synchronization pin is available for cases where a higher frequency (up to 500 kHz) is required. Pulse-by-pulse and frequency foldback overcurrent protection offer effective short circuit protection. Other features are voltage feed-forward, protection against feedback disconnection, inhibit and thermal shutdown. Figure 1. Demonstration board
L5970D (SO-8) board dimensions: 23 x 20 mm
Figure 2.
Package
Figure 3.
Pin connection
OUT SYNC INH
1 2 3 4
8 7 6 5
AM00004v1
VCC GND VREF FB
SO-8
COMP
May 2008
Rev 3
1/31
www.st.com
Contents
AN1330
Contents
1 Pin functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
1.1 1.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2
Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 Power supply and voltage reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Voltages monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Oscillator and synchronizer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 Current protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Error amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 PWM comparator and power stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 Inhibit function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 Thermal shutdown . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
3
Additional features and protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
3.1 3.2 3.3 Feedback disconnection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 Output overvoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 Zero load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
4
Closing the loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
4.1 4.2 4.3 Error amplifier and compensation network . . . . . . . . . . . . . . . . . . . . . . . . 12 LC filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 PWM comparator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
5
Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
5.1 5.2 5.3 5.4 5.5 Component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Short-circuit protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
6
Application ideas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
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AN1330
Contents
6.1 6.2 6.3 6.4
Positive buck-boost regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Buck-boost regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Dual output voltage with auxiliary winding . . . . . . . . . . . . . . . . . . . . . . . . 27 Synchronization example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
7
Compensation network with MLCC (multiple layer ceramic capacitor) at the output . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
7.1 External SOFT_START network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
8
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
3/31
List of figures
AN1330
List of figures
Figure 1. Figure 2. Figure 3. Figure 4. Figure 5. Figure 6. Figure 7. Figure 8. Figure 9. Figure 10. Figure 11. Figure 12. Figure 13. Figure 14. Figure 15. Figure 16. Figure 17. Figure 18. Figure 19. Figure 20. Figure 21. Figure 22. Figure 23. Figure 24. Figure 25. Figure 26. Figure 27. Figure 28. Figure 29. Figure 30. Demonstration board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Internal regulator circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 Oscillator circuit block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Current limitation circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Driving circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 Block diagram of the loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 Error amplifier equivalent circuit and compensation network . . . . . . . . . . . . . . . . . . . . . . . 13 Module plot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 Phase plot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 Layout example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Short-circuit current (VIN = 25 V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Short-circuit current (VIN = 30 V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Demonstration board application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 PCB layout (component side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 PCB layout (bottom side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 PCB layout (front side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Junction temperature vs. output current (VCC = 5 V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Junction temperature vs. output current (VCC = 12 V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Junction temperature vs. output current (VCC = 24 V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 Efficiency vs. output current (VCC = 5 V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 Junction temperature vs. output current (VCC = 12 V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 Positive buck-boost regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Buck-boost regulator. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Dual output voltage with auxiliary winding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 Synchronization example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 MLCC compensation network example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 Soft start network example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
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AN1330
Pin functions
1
1.1
Pin functions
Pin description
Table 1.
N. Name 1 OUT Regulator output
Pin description
Description
2
Master/slave synchronization. When open, a signal synchronous with the turn-OFF of the internal power is present. When connected to an external signal at a frequency SYNC higher than the internal one, then the device is synchronized by the external signal. Connecting the SYNC pins of two devices, the one with the higher frequency works as master and the other one works as slave. A logical signal (active high) disables the device. With an IHN higher than 2.2 V the device is OFF and with an INH lower than 0.8 V, the device is ON. If INH is not used the pin must be grounded. When it is open, an internal pull-up disables the device.
3
INH
4
COMP E/A output to be used for frequency compensation Step-down feedback input. Connecting the output voltage directly to this pin results in an output voltage of 1.235 V. An external resistor divider is required for higher output voltages (the typical value for the resistor connected between this pin and ground is 4.7 k). Reference voltage of 3.3 V. No filter capacitor is needed for stability Ground Unregulated DC input voltage.
5
FB
6 7 8
VREF GND VCC
1.2
Block diagram
Figure 4. Block diagram
VCC
TRIMMING
VOLTAGES MONITOR SUPPLY THERMAL SHUTDOWN
VREF BUFFER
VREF
1.235V 3.5V PEAK TO PEAK CURRENT LIMIT
INH COMP FB 1.235V SYNC
INHIBIT
E/A + + -
PWM
D
Q DRIVER FREQUENCY SHIFTER LPDMOS POWER
Ck
OSCILLATOR
GND
OUT
AM00003v1
5/31
Functional description
AN1330
2
Functional description
The main internal blocks are shown in the device block diagram in Figure 4. They are:

A voltage regulator supplying the internal circuitry. From this regulator, a 3.3 V reference voltage is externally available. A voltage monitor circuit which checks the input and internal voltages. A fully integrated sawtooth oscillator with a frequency of 250 kHz 15%, including also the voltage feed forward function and an input/output synchronization pin. Two embedded current limitation circuits which control the current that flows through the power switch. The pulse-by-pulse current limit forces the power switch OFF cycle by cycle if the current reaches an internal threshold, while the frequency shifter reduces the switching frequency in order to significantly reduce the duty cycle. A transconductance error amplifier. A pulse width modulation (PWM) comparator and the relative logic circuitry necessary to drive the internal power. A high side driver for the internal P-MOS switch. An inhibit block for standby operation. A circuit to implement the thermal protection function.

2.1
Power supply and voltage reference
The internal regulator circuit (shown in Figure 5) consists of a start-up circuit, an internal voltage Preregulator, the Bandgap voltage reference and the Bias block that provides current to all the blocks. The Starter gives the start-up currents to the entire device when the input voltage goes high and the device is enabled (inhibit pin connected to ground). The Preregulator block supplies the Bandgap cell with a preregulated voltage VREG that has a very low supply voltage noise sensitivity.
2.2
Voltages monitor
An internal block continuously senses the VCC, VREF and VBG. If the voltages go higher than their thresholds, the regulator begins operating. There is also a hysteresis on the VCC (UVLO).
6/31
AN1330 Figure 5. Internal regulator circuit
VCC
Functional description
STARTER
PREREGULATOR VREG BANDGAP
IC BIAS
VREF
AM00006v1
2.3
Oscillator and synchronizer
Figure 6 shows the block diagram of the oscillator circuit. The Clock Generator provides the switching frequency of the device, which is internally fixed at 250 kHz. The Frequency Shifter block acts to reduce the switching frequency in case of strong overcurrent or short circuit. The clock signal is then used in the internal logic circuitry and is the input of the Ramp Generator and synchronizer blocks. The Ramp Generator circuit provides the sawtooth signal, used for PWM control and the internal voltage feed-forward, while the synchronization circuit generates the synchronization signal. The device also has a synchronization pin which can works both as Master and Slave. As Master, it serves to synchronize external devices to the internal switching frequency, and as Slave to synchronize itself using an external signal up to 500 kHz. In particular, when connecting together two devices the one with the lower switching frequency works as Slave and the other as Master. To synchronize the device, the SYNC pin must pass from a low level to a level higher than the synchronization threshold with a duty cycle that can vary from approximately 10% to 90%, depending also on the signal frequency and amplitude. The frequency of the synchronization signal must be, at a minimum, higher than the internal switching frequency of the device (250 kHz).
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Functional description Figure 6. Oscillator circuit block diagram
FREQUENCY FREQUENCY SHIFTER SHIFTER
AN1330
CLOCK
t
Ibias_osc
CLOCK CLOCK GENERATOR GENERATOR RAMP RAMP GENERATOR GENERATOR
RAMP
SYNCHRONIZATOR SYNCHRONIZER SYNC
AM00007v1
2.4
Current protection
The L5970D has two types of current limit protection: pulse-by-pulse and frequency foldback. The schematic of the current limitation circuitry for the pulse-by-pulse protection is shown in Figure 7. The output power PDMOS transistor is split into two parallel PDMOS transistors. The smallest one includes a resistor in series, RSENSE. The current is sensed through RSENSE and if reaches the threshold, the mirror becomes unbalanced and the PDMOS is switched off until the next falling edge of the internal clock pulse. Due to this reduction of the ON time, the output voltage decreases. Since the minimum switch ON time (necessary to avoid a false overcurrent signal) is too short to obtain a sufficiently low duty cycle at 250 kHz, the output current, in strong overcurrent or short circuit conditions, could increase again. For this reason the switching frequency is also reduced, thus keeping the inductor current under its maximum threshold. The Frequency Shifter (Figure 6) functions based on the feedback voltage. As the feedback voltage decreases (due to the reduced duty cycle), the switching frequency decreases also. Figure 7. Current limitation circuitry
VCC IOFF DRIVER A1 A2 IL RSENSE RTH
OUT A1/A2=95 I PWM
AM00008v1
I
NOT
2.5
Error amplifier
The voltage error amplifier is the core of the loop regulation. It is a transconductance operational amplifier whose non inverting input is connected to the internal voltage
8/31
AN1330
Functional description reference (1.235 V), while the inverting input (FB) is connected to the external divider or directly to the output voltage. The output (COMP) is connected to the external compensation network. The uncompensated error amplifier has the following characteristics: Table 2. Uncompensated error amplifier characteristics
Description Transconductance Low frequency gain Minimum sink/source voltage Output voltage swing Input bias current Values 2300 S 65 dB 1500 A/300 A 0.4 V/3.65 V 2.5 A
The error amplifier output is compared with the oscillator sawtooth to perform PWM control.
2.6
PWM comparator and power stage
This block compares the oscillator sawtooth and the error amplifier output signals generating the PWM signal for the driving stage. The power stage is a highly critical block, as it functions to guarantee a correct turn ON and turn OFF of the PDMOS. The turn ON of the power element, or more accurately, the rise time of the current at turn ON, is a very critical parameter. At a first approach, it appears that the faster the rise time, the lower the turn ON losses. However, there is a limit introduced by the recovery time of the recirculation diode. In fact, when the current of the power element is equal to the inductor current, the diode turns OFF and the drain of the power is able to go high. But during its recovery time, the diode can be considered a high value capacitor and this produces a very high peak current, responsible of many problems:

Spikes on the device supply voltage that cause oscillations (and thus noise) due to the board parasitics Turn ON overcurrent leads to a decrease in the efficiency and system reliability Major EMI problems Shorter freewheeling diode life
The fall time of the current during the turn OFF is also critical, as it produces voltage spikes (due to the parasitic elements of the board) that increase the voltage drop across the PDMOS. In order to minimize these problems, a new driving circuit topology has been used and the block diagram is shown in Figure 8. The basic idea is to change the current levels used to turn the power switch ON and OFF, based on the PDMOS and the gate clamp status. This circuitry allows the power switch to be turned OFF and ON quickly and addresses the freewheeling diode recovery time problem. The gate clamp is necessary to avoid that VGS of the internal switch goes higher than VGSmax. The ON/OFF Control block protects against any cross conduction between the supply line and ground.
9/31
Functional description Figure 8. Driving circuitry
VCC
AN1330
Vgsmax IOFF CLAMP GATE
PDMOS DRAIN VOUT L ESR ILOAD
STOP DRIVE DRAIN ON/OFF CONTROL
OFF
ON C ION
AM00009v1
2.7
Inhibit function
The inhibit feature is used to put the device in standby mode. With the INH pin higher than 2.2 V the device is disabled and the power consumption is reduced to less than 100 A. With the INH pin lower than 0.8 V, the device is enabled. If the INH pin is left floating, an internal pull up ensures that the voltage at the pin reaches the inhibit threshold and the device is disabled. The pin is also VCC compatible.
2.8
Thermal shutdown
The shutdown block generates a signal that turns OFF the power stage if the temperature of the chip goes higher than a fixed internal threshold (150 C). The sensing element of the chip is very close to the PDMOS area, ensuring fast and accurate temperature detection. A hysteresis of approximately 20 C avoids that the devices turns ON and OFF continuously.
10/31
AN1330
Additional features and protection
3
3.1
Additional features and protection
Feedback disconnection
If the feedback is disconnected, the duty cycle increases towards the maximum allowed value, bringing the output voltage close to the input supply. This condition could destroy the load. To avoid this hazardous condition, the device is turned OFF if the feedback pin is left floating.
3.2
Output overvoltage protection
Overvoltage protection, or OVP, is achieved by using an internal comparator connected to the feedback, which turns OFF the power stage when the OVP threshold is reached. This threshold is typically 30% higher than the feedback voltage. When a voltage divider is required to adjust the output voltage (Figure 14), the OVP intervention will be set at: Equation 1 V OVP = 1.3 * --------------------- * V FB R2 R1 + R2
Where R1 is the resistor connected between the output voltage and the feedback pin, and R2 is between the feedback pin and ground.
3.3
Zero load
Due to the fact that the internal power is a PDMOS, no bootstrap capacitor is required and so the device works properly even with no load at the output. In this condition it works in burst mode, with random burst repetition rate.
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Closing the loop
AN1330
4
Closing the loop
Figure 9. Block diagram of the loop
4.1
Error amplifier and compensation network
The output L-C filter of a step-down converter contributes with 180 degrees phase shift in the control loop. For this reason a compensation network between the COMP pin and GROUND is added. The simplest compensation network together with the equivalent circuit of the error amplifier are shown in Figure 10. RC and CC introduce a pole and a zero in the open loop gain. CP does not significantly affect system stability but it is useful to reduce the noise of the COMP pin. The transfer function of the error amplifier and its compensation network is: Equation 2
A V0 * ( 1 + s * R c * C c ) A 0 ( s ) = --------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------2 s * R0 * ( C0 + Cp ) * Rc * Cc + s * ( R0 * Cc + R0 * ( C0 + Cp ) + Rc * Cc ) + 1
Where Avo = Gm * Ro
12/31
AN1330
Closing the loop Figure 10. Error amplifier equivalent circuit and compensation network
The poles of this transfer function are (if Cc >> C0+CP): Equation 3
1 F P1 = ------------------------------------2 * * R0 * Cc
Equation 4
1 F P2 = ------------------------------------------------------2 * * Rc * ( C0 + Cp )
where the zero is defined as: Equation 5
1 F Z1 = -----------------------------------2 * * Rc * Cc
FP1 is the low frequency which sets the bandwidth, while the zero FZ1 is usually put near to the frequency of the double pole of the L-C filter (see below). FP2 is usually at a very high frequency.
4.2
LC filter
The transfer function of the L-C filter is given by: Equation 6
R LOAD * ( 1 + ESR * C OUT * s ) A LC ( s ) = ---------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------2 s * L * C OUT * ( ESR + R LOAD ) + s * ( ESR * C OUT * R LOAD + L ) + R LOAD
where RLOAD is defined as the ratio between VOUT and IOUT. If RLOAD>>ESR, the previous expression of ALC can be simplified and becomes:
13/31
Closing the loop Equation 7
1 + ESR * C OUT * s A LC ( s ) = --------------------------------------------------------------------------------------------2 L * C OUT * s + ESR * C OUT * s + 1
AN1330
The zero of this transfer function is given by: Equation 8
1 F O = --------------------------------------------------2 * * ESR * C OUT
F0 is the zero introduced by the ESR of the output capacitor and it is very important to increase the phase margin of the loop. The poles of the transfer function can be calculated through the following expression: Equation 9
2 - ESR * C OUT ( ESR * C OUT ) - 4 * L * C OUT F PLC1, 2 = -----------------------------------------------------------------------------------------------------------------------------------------2 * L * C OUT
In the denominator of ALC the typical second order system equation can be recognized: Equation 10
s + 2 * * n * s +
2 2 n
If the damping coefficient is very close to zero, the roots of the equation become a double root whose value is n. Similarly, for ALC the poles can usually be defined as a double pole whose value is: Equation 11
1 F PLC = --------------------------------------------2 * * L * C OUT
4.3
PWM comparator
The PWM gain is given by the following formula: Equation 12
V cc G PWM ( s ) = ------------------------------------------------------------( V OSCMAX - V OSCMIN )
where VOSCMAX is the maximum value of a sawtooth waveform and VOSCMIN is the minimum value. A voltage feed forward is implemented to ensure a constant GPWM. This is obtained by generating a sawtooth waveform directly proportional to the input voltage VCC. Equation 13
V OSCMAX - V OSCMIN = K * V CC
Where K is equal to 0.076. Therefore the PWM gain is also equal to: Equation 14
1 G PWM ( s ) = --- = const K
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AN1330
Closing the loop This means that even if the input voltage changes, the error amplifier does not change its value to keep the loop in regulation, thus ensuring a better line regulation and line transient response. To sum up the Open Loop Gain can be written as: Equation 15
R2 G ( s ) = G PWM ( s ) * ------------------- * A O ( s ) * A LC ( s ) R1 + R2
Example: Considering RC = 2.7 k, CC = 22 nF and CP = 220 pF, the poles and zeroes of A0 are: FP1 = 9 Hz FP2 = 256 kHz FZ1 = 2.68 kHz If L = 22 H, COUT = 100 F and ESR = 80 m, the poles and zeroes of ALC become: FPLC = 3.39 kHz F0 = 19.89 kHz Finally R1 = 5.6 k and R2 = 3.3 k. The gain and phase bode diagrams are plotted respectively in Figure 11 and Figure 12. Figure 11. Module plot
15/31
Closing the loop Figure 12. Phase plot
AN1330
The cut-off frequency and the phase margin are: Equation 16
F C = 22.8kHz Phase margin = 39.8
16/31
AN1330
Application information
5
5.1
Application information
Component selection
Input capacitor
The input capacitor must be able to withstand the maximum input operating voltage and the maximum RMS input current. Since step-down converters draw current from the input in pulses, the input current is squared and the height of each pulse is equal to the output current. The input capacitor has to absorb all this switching current, which can be up to the load current divided by two (worst case, with duty cycle of 50%). For this reason, the quality of these capacitors has to be very high to minimize the power dissipation generated by the internal ESR, thereby improving system reliability and efficiency. The critical parameter is usually the RMS current rating, which must be higher than the RMS input current. The maximum RMS input current (flowing through the input capacitor) is: Equation 17 I RMS = I O 2*D * D - ----------------- + D - ------ 2 2
Where is the expected system efficiency, D is the duty cycle and IO the output DC current. This function reaches its maximum value at D = 0.5 and the equivalent RMS current is equal to IO divided by 2 (considering = 1). The maximum and minimum duty cycles are: Equation 18 V OUT + V F D MAX = --------------------------------------V INMIN - V SW and V OUT + V F D MIN = ----------------------------------------V INMAX - V SW
Where VF is the freewheeling diode forward voltage and VSW the voltage drop across the internal PDMOS. Considering the range DMIN to DMAX, it is possible to determine the max IRMS going through the input capacitor. Capacitors that can be considered are: - Electrolytic capacitors: These are widely used due to their low price and their availability in a wide range of RMS current ratings. The only drawback is that, considering ripple current rating requirements, they are physically larger than other capacitors. Ceramic capacitors: If available for the required value and voltage rating, these capacitors usually have a higher RMS current rating for a given physical dimension (due to very low ESR). The drawback is the considerably high cost. Tantalum capacitors: Good, small tantalum capacitors with very low ESR are becoming more available. However, they can occasionally burn if subjected to very high current during charge. Therefore, it is better to avoid this type of capacitor for the input filter of the device. They can, however, be subjected to high surge current when connected to the power supply.
-
-
Output capacitor
The output capacitor is very important to meet the output voltage ripple requirement. Using a small inductor value is useful to reduce the size of the choke but it increases the current ripple. So, to reduce the output voltage ripple, a low ESR capacitor is required.
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Application information
AN1330
Nevertheless, the ESR of the output capacitor introduces a zero in the open loop gain, which helps to increase the phase margin of the system. If the zero goes to a very high frequency, its effect is negligible. For this reason, ceramic capacitors and very low ESR capacitors in general should be avoided. Tantalum and electrolytic capacitors are usually a good choice for this purpose. Table 3 below provides a list of some tantalum capacitor manufacturers. Table 3. Output capacitor selection
Series TPS T494/5
(1)
Manufacturer AVX KEMET Sanyo POSCAP Sprague
Cap value (F) 100 to 470 100 to 470 100 to 470 220 to 390
Rated voltage (V) 4 to 35 4 to 20 4 to 16 4 to 20
ESR (m) 50 to 200 30 to 200 40 to 80 160 to 650
TPA/B/C 595D
1. POSCAP capacitors have characteristic very similar to tantalum capacitors.
Inductor
The inductor value is very important because it fixes the ripple current flowing through output capacitor. The ripple current is usually fixed at 20-40% of IOmax, which is 0.2 - 0.4 A with IOmax = 1 A. The approximate inductor value is obtained using the following formula: Equation 19 ( V IN - V OUT ) L = -------------------------------------I
* T ON
where TON is the ON time of the internal switch, given by D * T. For example, with VOUT = 3.3 V, VIN = 12 V and IO = 0.3 A, the inductor value is about 35 H. The peak current through the inductor is given by: Equation 20 I I PK = I O + ---2 and it can be observed that if the inductor value decreases, the peak current (which must be lower than the current limit of the device) increases. So, when the peak current is fixed, a higher inductor value allows a higher value for the output current. In Table 4: Inductor selection, some inductor manufacturers are listed. Table 4. Inductor selection
Series DO1813HC Coilcraft DO3316 33 to 47 1.6 to 2 Inductor value (H) 22 to 33 Saturation current (A) 1 to 1.2
Manufacturer
18/31
AN1330 Table 4. Inductor selection (continued)
Series UP1B Coiltronics UP2B HM76-2 BI HM76-3 Murata Panasonic Sumida Epcos Wurth Elektronik LQN6C ELLATV CR75 B82476 744561 33 to 47 22 to 33 22 to 47 22 to 33 33 to 47 33 to 47 33 to 47 22 to 33
Application information
Manufacturer
Inductor value (H) 22 to 33
Saturation current (A) 1 to 1.2 1.7 to 2 1 to 1.2 2 to 2.5 0.9 to 1.2 1.4 to 2.05 1.2 to 1.5 1.6 to 2 1.6 to 2
5.2
Layout considerations
The layout of switching DC-DC converters is very important to minimize noise and interference. Power-generating portions of the layout are the main cause of noise and so high switching current loop areas should be kept as small as possible and lead lengths as short as possible. High impedance paths (in particular the feedback connections) are susceptible to interference, so they should be as far as possible from the high current paths. A layout example is provided in Figure 13 below. The input and output loops are minimized to avoid radiation and high frequency resonance problems. The feedback pin connections to the external divider are very close to the device to avoid pick-up noise. Moreover the GND pin of the device is connected to the ground plane directly with VIA on the bottom side of the PCB. Figure 13. Layout example
COMPENSATION NETWORK FAR FROM HIGH CURRENT PATHS
to output voltage
MINIMUN SIZE OF FEEDBACK PIN CONNECTIONS TO AVOID PICKUP
Inhibit signal
R2 5 R1 L5970 8
CONNECTION TO GROUNDPLANE THROUGH VIA
4 L 1 Vout
Vin Cin D Cout
Gnd
VERY SMALL HIGH CURRENT CIRCULATING PATH TO MINIMIZE RADIATION AND HIGH FREQUENCY RESONANCE PROBLEMS OUTPUT CAPACITOR DIRECTLY CONNECTED TO HEAVY GROUND
AM00012v1
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Application information
AN1330
5.3
Thermal considerations
The dissipated power of the device is tied to three different sources:
switching losses due to the not negligible RDSON. These are equal to:
Equation 21
2 P ON = R DS ( on ) * ( I OUT ) * D
Where D is the duty cycle of the application. Note that the duty cycle is theoretically given by the ratio between VOUT and VIN, but in practice it is substantially higher than this value to compensate for the losses of the overall application. For this reason, the switching losses related to the RDSON increase compared to an ideal case.
Switching losses due to turning ON and OFF. These are derived using the following equation:
Equation 22
( T ON + T OFF ) P SW = V IN * I OUT * ---------------------------------------- * F SW = V IN * I OUT * T SW * F SW 2
where TON and TOFF are the overlap times of the voltage across the power switch and the current flowing into it during the turn ON and turn OFF phases. TSW is the equivalent switching time.
Quiescent current losses.
Equation 23
P Q = V IN * I Q
Where IQ is the quiescent current.
Example: - - - VIN = 5 V VOUT = 3.3 V IOUT = 1 A
RDSON has a typical value of 0.25 @ 25 C and increases up to a maximum value of 0.5 @ 150 C. We can consider a value of 0.4 . TSW is approximately 120 ns. IQ has a typical value of 2.5 mA @ VIN = 12 V. The overall losses are: Equation 24
2 P TOT = R DSON ( I OUT ) D + V IN I OUT T SW F SW + V IN I Q = 2 -9 3 -3 0.4 1 0.7 + 5 1 120 10 250 10 + 5 2.5 10 0.44W
The junction temperature of the device will be: Equation 25
T J = T A + Rth J - A * P TOT
where TA is the ambient temperature and RthJ-A is the thermal resistance junction-toambient.
20/31
AN1330
Application information Considering the device in an SO-8 package mounted on the board with a good groundplane, that it has a thermal resistance-junction to-ambient (RthJ-A) of about 115 C/W and an ambient temperature of about 70 C. Equation 26
T J = 70 + 0.44 * 115 121C
5.4
Short-circuit protection
In overcurrent protection mode, when the peak current reaches the current limit, the device reduces the TON down to its minimum value (approximately 250 ns) and the switching frequency to approximately one third of its nominal value (see Section 2.4: Current protection). In these conditions, the duty cycle is strongly reduced and, in most of applications, this is enough to limit the current to ILIM. In any event, in case of heavy shortcircuit at the output (VOUT=0 V) and depending on the application conditions (VCC value and parasitic effect of external components) the current peak could reach values higher than ILIM. This can be understood considering the inductor current ripple during the ON and OFF phases:
ON phase
Equation 27
( VIN - Vout - DCRL * I ) I L = -------------------------------------------------------------------L
* TON
OFF phase
( V D + V out + DCR L * I ) I L = ----------------------------------------------------------------L
Equation 28
* T OFF
where VD is the voltage drop across the diode, and DCRL is the series resistance of the inductor. In short-circuit conditions, VOUT is negligible. So, during the TOFF, the voltage applied to the inductor is very small and it may be that the current ripple in this phase does not compensate for the current ripple during the TON. The maximum current peak can be easily measured through the inductor with VOUT=0 V (short-circuit) and VCC=VINmax. In cases where the application must sustain the shortcircuit condition for an extended period, the external components (mainly the inductor and diode) must be selected based on this value.
21/31
Application information Figure 14. Short-circuit current (VIN = 25 V)
AN1330
Figure 15. Short-circuit current (VIN = 30 V)
In Figure 14 and Figure 15, for example, it can be observed that when the input voltage increases for a given component list, the current peak increases also. The current limit is immediately triggered but the current peak increases until the current ripple during the TOFF is equal to the current ripple during the TON.
5.5
Application circuit
Figure 16 shows the demonstration board application circuit for the device in the SMD version, where the input supply voltage, VCC, can range from 4.4 V to 25 V due to the rated voltage of the input capacitor and the output voltage is adjustable from 1.235 V to VCC.
22/31
AN1330 Figure 16. Demonstration board application circuit
3.3V VIN = 4.4V to 25V VREF VCC SYNC. C1 10F 25V CERAMIC COMP C4 22nF C3 220pF 6 8 2 4 INH. R3 4.7K 1 OUT L1 33H
Application information
VOUT=3.3V
L5970D
3 7 5 GND FB
D1 STPS2L25U
R1 5.6K
C2 100F 10V
R2 3.3K
AM00017v1
Table 5.
Component list
Part number GRM32DR61E106KA12L POSCAP 10TPB100M C1206C221J5GAC C1206C223K5RAC Description 10 F, 25 V 100 F, 10 V 220 pF, 5%, 50 V 22 nF, 10%, 50 V 5.6 K, 1%, 0.1 W 0603 3.3 K, 1%, 0.1 W 0603 4.7 K, 1%, 0.1 W 0603 STPS2L25U DO3316P-333 2 A, 25 V 33 H, 2 A Manufacturer Murata Sanyo KEMET KEMET Neohm Neohm Neohm STMicroelectronics Coilcraft
Reference C1 C2 C3 C4 R1 R2 R3 D1 L1
Figure 17. PCB layout (component side)
L5970D
EVAL.BOARD R2 REF R1 C3 VIN U1 VOUT C1 GND
www.st.com
AM00018v1
R INH L1 SYN
R3 C4
D1 C2 GND.
23/31
Application information Figure 18. PCB layout (bottom side)
AN1330
AM00019v1
Figure 19. PCB layout (front side)
AM00020v1
Below, some graphs are provided which show the Tj versus output current in different input and output voltage conditions, as well as some efficiency measurements.
Figure 20. Junction temperature vs. output current (VCC = 5 V)
Tj(C)
Figure 21. Junction temperature vs. output current (VCC = 12 V)
Tj(C)
130 120 110 100 90 80 70 60 50 40 30 20 0.2
Vo=3.3V
Vo=2.5V
Vcc=5V Tamb=25C
Vo=1.8V
130 120 110 100 90 80 70 60 50 40 30 20 0.2
Vo=5V
Vo=3.3V
Vcc=12V Tamb=25C
Vo=2.5V
0.4
0.6
0.8
Io(A)
1
1.2
1.4
1.6
0.4
0.6
0.8
Io(A)
1
1.2
1.4
1.6
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AN1330
Application information
Figure 22. Junction temperature vs. output current (VCC = 24 V)
Tj(C)
Figure 23. Efficiency vs. output current (VCC = 5 V)
94 92 90 88 86 84 82 80 78 76 74 72 70 0.1
140
Vo=18V
Vo=12V
100 80 60 40 20 0 0.2
Vcc=24V Tamb=25C
Vo=5V
Efficiency (%)
120
Vo=3.3V Vo=2.5V
Vo=1.8V
Vcc=5V 0.2 0.3 0.4 0.5 0.6 Io (A) 0.7 0.8 0.9 1
0.4
0.6
0.8
Io(A)
1
1.2
1.4
Figure 24. Junction temperature vs. output current (VCC = 12 V)
92 90 88 86 84 82 80 78 76 74 72 70 Vo=5V
Efficiency (%)
Vo=3.3V
Vo=2.5V Vcc=12V
0.1
0.2
0.3
0.4
0.5 0.6 Io (A)
0.7
0.8
0.9
1
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Application ideas
AN1330
6
6.1
Application ideas
Positive buck-boost regulator
The device can be used to implement a step-up/down converter with a positive output voltage. Figure 25 below shows the schematic diagram of this topology for an output voltage of 12 V. The input voltage can range from 5 V and 35 V. The output voltage is given by VO=VIN * D/(1-D), where D is duty cycle. The maximum output current is given by IOUT=1x (1-D). The current capability is reduced by the term (1-D) and so, for example, with a duty cycle of 0.5, the maximum output current deliverable to the load is 0.5 A. This is due to the fact that the current flowing trough the internal power switch is delivered to the output only during the OFF phase. Figure 25. Positive buck-boost regulator
VIN=5V
L1 33uH
VCC
8 U1 L5970 4 2 6
VREF GND
OUT
D2 STPS2L25U
VOUT=12V/0.3A
1
D1 STPS2L25U 24k M1 STN4NE03L 2.7k C4 100uF 16V
COMP C1 10uF 25V Ceramic
7
5 3
FB
C2 220pF
C3 22nF R3 4.7k
SYNC
INH
3.3V
Vin=5V Vout=12V
Iout=0.3A
Efficiency=76%
6.2
Buck-boost regulator
In Figure 26, the schematic circuit for a standard buck-boost topology is shown. The output voltage is given by VO = - VIN * D/(1-D). The maximum output current is equal to IOUT=1 * (1D), for the same reason as that of the up-down converter. An important thing to take into account is that the ground pin of the device is connected to the negative output voltage. Therefore, the device is subjected to a voltage equal to VIN-VO, that has to be lower than 36 V (the maximum operating input voltage). Figure 26. Buck-boost regulator
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AN1330
Application ideas
6.3
Dual output voltage with auxiliary winding
When two output voltages are required, it is possible to create a dual output voltage converter by using a coupled inductor. During the ON phase, the current is delivered to VOUT while D2 is reverse-biased. During the OFF phase, the current is delivered through the auxiliary winding to the output voltage VOUT1. This is possible only if the magnetic core has stored sufficient energy. So, to be certain that the application is working properly, the load related to the second output VOUT1 should be much lower than the load related to VOUT. Figure 27. Dual output voltage with auxiliary winding
6.4
Synchronization example
Two or more devices (up to 6) can be synchronized simply by connecting the synchronization pins. In this case, the device with a slightly higher switching frequency value will work as master and the ones with a slightly lower switching frequency values will work as a slaves. The device can also be synchronized from an external source. In this case the logic signal must have a frequency higher than the internal switching frequency of the device (250 kHz). Figure 28. Synchronization example
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Compensation network with MLCC (multiple layer ceramic capacitor) at the output
AN1330
7
Compensation network with MLCC (multiple layer ceramic capacitor) at the output
MLCCs with values in the range of 10 F-22 F and rated voltages in the range of 10 V-25 V are available today at relatively low cost from many manufacturers. These capacitors have very low ESR values (a few m) and thus are occasionally used for the output filter in order to reduce the voltage ripple and the overall size of the application. However, a very low ESR value affects the compensation of the loop (see Section 4: Closing the loop) and in order to keep the system stable, a more complicated compensation network may be required. Figure 29 shows an example of a compensation network that stabilizes the system with ceramic capacitors at the output (the optimum component value depends on the application). Figure 29. MLCC compensation network example
7.1
External SOFT_START network
At the start-up, the device can quickly increase the current up to the current limit in order to charge the output capacitor. If a soft ramp-up of the output voltage is required, an external soft-start network can be implemented as shown in Figure 30. The capacitor C is charged up to an external reference (through R), and the BJT clamps the COMP pin. This clamps the duty cycle, limiting the slew rate of the output voltage.
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AN1330
Compensation network with MLCC (multiple layer ceramic capacitor) at the output Figure 30. Soft start network example
VIN=4.4V to 25V VREF
R=4K7
Vcc
OUT
33uH Coilcraft
D1
VOUT=3.3V
8
VREF COMP C1 10uF 25V CERAMIC
BC327 Css=2.7nF C4=22nF
1 6 L5970D 7
GND FB
L1
STPS2L25U
R1=5.6K
4 2
5 3
INH
R2=3.3K
C2 100uF 10V
SYNC
C3=220pF R3=4.7K
AM00026v1
29/31
Revision history
AN1330
8
Revision history
Table 6.
Date
Document revision history
Revision Changes - the document has been reformatted - Section 4: Closing the loop modified - Minor text changes - Restructured document sections - Minor text changes
22-May-2007
2
19-May-2008
3
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AN1330
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